Front end sample and hold circuit for a reconfigurable analog-to-digital converter

ABSTRACT

A reconfigurable wideband analog-to-digital converter (ADC) system comprising a first converter stage including: a first signal path including a first sample and hold circuit responsive to an input signal for sampling the input signal at a first resolution; a second signal path arranged parallel to the first signal path and including a second sample and hold circuit responsive to the input signal for sampling the input signal at a second resolution; and a control circuit arranged in series with the first and second signal paths and configured to isolate one of the first and second signal paths from an output of the control circuit in response to a control signal.

FIELD OF THE INVENTION

The present disclosure relates generally to analog to digitalconverters, and more specifically, to sample and hold circuits foranalog to digital converter topologies.

BACKGROUND

An analog-to-digital converter (A/D or ADC) is an electronic device thatcan convert analog signal information (e.g. amplitude or phase) to adigital (e.g. numerical) value representative of the analog signal.These devices enable a central processing unit to carry out processingfunctions in a more quantized mathematical domain without the need fordownstream analog devices.

Signal processing systems utilizing A/D converters, including complexA/D converters (CADCs), have various functionality requirements. Suchrequirements may include a track and hold, or sample and hold operationperformed on an analog input signal, as well as precise timingoperations, in addition to the A/D conversion process. Clock generators,for example, are used to synchronize sample and hold amplifiers with anADC. The sample and hold amplifiers follow an input analog signal ofinterest until a control signal from the clock causes the amplifier tofreeze and hold the time varying analog signal (for a given timeinterval). The same clock signal also strobes the ADC to convert theheld or “frozen” sample to a digital value. This digital data can thenbe buffered and read out to memory for further processing. The time ittakes for the sample and hold amplifiers and ADC to perform theoperation and be ready for the next value is the A/D sample rate.

Standard A/D conversion systems are too slow to directly digitize ultrahigh frequency and microwave RF signals. These frequency ranges aretypically in the second and third Nyquist regions of the converter,including military frequency bands up to 20-40 GHz. Accordingly, thesesystems must utilize several analog down conversion steps before thesignal is sufficiently low in frequency to allow for digitization. Inaddition to limited or low sample rates, A/D converters further hindersignal processing operations due to their limited resolution, measuredby the effective number of bits or ENOB. In order to improve ADCresolution, various architectures including pipeline ADCs have beendeveloped. Digitally programmable ADCs have been developed to supportcertain processing requirements such as multi-band and multi-modeoperation. One such example is a digitally programmable sub-ranging ADC,which incorporates the improved resolution of pipeline-basedarchitectures with the ability to operate in various modes (e.g. 5 bit,9 bit and 13 bit). However, these systems typically incorporate analogmultiplexers MUXs in a primary analog signal path of sub-ranging ADCsystem in order to reconfigure the system for different operatingresolutions. A drawback of this approach is that these MUXs must possessa dynamic range greater than or equal to that of the converter as awhole. Further, because the MUXs are always in the primary signal path,they will necessarily add noise, and degrade settling and dynamic rangeof the converter.

Alternative converter topologies are desired which accommodatemulti-band and multi-mode operation, in addition to being digitallyprogrammable, and preferably without one or more of the above-describedlimitations.

SUMMARY

In one embodiment, a reconfigurable wideband analog-to-digital converter(ADC) system is provided. The system includes a first converter stagedefining a first signal path having a first sample and hold circuitresponsive to an input signal for sampling the input signal at a firstresolution, and a second signal path arranged parallel to the firstsignal path and including a second sample and hold circuit responsive tothe input signal for sampling the input signal at a second resolution. Acontrol circuit is arranged in series with the first and second signalpaths and configured to isolate one of the first and second signal pathsfrom an output of the control circuit in response to a control signalsupplied to at least one transistor. In one embodiment, the controlcircuit comprises a first transistor having a base terminal connected toan output of the first signal path and an emitter terminal connected tothe output of the control circuit, and a second transistor having a baseterminal connected to an output of the second signal path and an emitterterminal connected to the output of the control circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified schematic diagram of a sub-ranging ADC accordingto the prior art.

FIG. 2 is a simplified schematic diagram illustrating a clock generationscheme for the sub-ranging ADC of FIG. 1.

FIG. 3 is a simplified schematic diagram of a digitally programmablesub-ranging ADC according to the prior art.

FIG. 4 a is a simplified schematic diagram of a digitally programmablesub-ranging ADC according to embodiments of the present disclosure.

FIG. 4 b is a section view of a portion of the digitally programmablesub-ranging ADC of FIG. 4 a.

FIG. 5 is a simplified schematic diagram of a unit delay-locked loop(DLL) used in a clock generation system according to embodiments of thepresent disclosure.

FIG. 6 is a simplified schematic diagram of a clock generation systemutilizing a plurality of DLLs, as illustrated in FIG. 5, according to anembodiment of the present disclosure.

FIG. 7 is a simplified schematic diagram of a clock edge control systemfor controlling DACs shown in FIG. 6 according to an embodiment of thepresent disclosure.

FIG. 8 a is a simplified schematic diagram of a reconfigurablesub-ranging ADC according to an embodiment of the present disclosure.

FIG. 8 b is a simplified diagram of a reconfigurable ADC and DAC used inthe sub-ranging ADC of FIG. 8 a.

FIG. 9 is a simplified schematic diagram of a reconfigurable DAC bufferaccording to an embodiment of the present disclosure.

FIG. 10 is a graphical illustration of the operation of a reconfigurableADC according to an embodiment of the present disclosure.

FIGS. 11 a and 11 b are simplified schematic diagrams of areconfigurable ADC according to an embodiment of the present disclosure.

FIG. 12 is a simplified schematic diagram of a reconfigurable DACaccording to an embodiment of the present disclosure.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

It is to be understood that the figures and descriptions of the presentinvention have been simplified to illustrate elements that are relevantfor a clear understanding of the present invention, while eliminating,for purposes of clarity, many other elements found in ADCs and theirassociated clock generation systems, including digitally programmablesub-ranging ADCs. However, because such elements are well known in theart, and because they do not facilitate a better understanding of thepresent invention, a discussion of such elements is not provided herein.The disclosure herein is directed to all such variations andmodifications known to those skilled in the art.

In the following detailed description, reference is made to theaccompanying drawings that show, by way of illustration, specificembodiments in which the invention may be practiced. It is to beunderstood that the various embodiments of the invention, althoughdifferent, are not necessarily mutually exclusive. Furthermore, aparticular feature, structure, or characteristic described herein inconnection with one embodiment may be implemented within otherembodiments without departing from the scope of the invention. Inaddition, it is to be understood that the location or arrangement ofindividual elements within each disclosed embodiment may be modifiedwithout departing from the scope of the invention. The followingdetailed description is, therefore, not to be taken in a limiting sense,and the scope of the present invention is defined only by the appendedclaims, appropriately interpreted, along with the full range ofequivalents to which the claims are entitled. In the drawings, likenumerals refer to the same or similar functionality throughout severalviews.

Embodiments of the present disclosure include improved ADC architecturesthat support multiple operational bandwidths as well as multiple modesof operation. This ADC architecture is based on sub-ranging ADCarchitectures, which have historically addressed both narrowband, highdynamic range applications for radar, as well as moderate bandwidth forcommunications applications.

FIG. 1 shows a simplified diagram of an exemplary sub-ranging ADC system10 according to the prior art with three (3) sampling passes: first pass(coarse) 12, second pass (fine) 14, and third pass (superfine) 16. TheADC operates in a conventional way, with an analog input signal 11sampled and held by a sample and hold circuit comprising a first bufferamplifier 13 and a first hold amplifier 15, while a first ADC 17quantizes a first number of bits. The output of ADC 17 is provided to afirst digital-to-analog converter (DAC) 18. The analog output of DAC 18is subtracted from the output of a second sample and hold circuit,including a second buffer amplifier 13′ and second hold amplifier 15′via amplifier 19. The resulting signal is provided to a third sample andhold circuit of second pass 14, comprising a third buffer amplifier 13″and a third hold amplifier 15″. Second sampling pass 14 further includesa second ADC 17′ and DAC 18′. The output of DAC 18′ is again subtractedfrom the input signal via amplifier 19′, and buffered to a third ADC 17″which constitutes third pass or stage 16 of ADC system 10. When a stagefinishes processing a sample, determining bits, and passing the residueto the next stage, processing the next sample received from thesample-and-hold embedded within each stage may begin.

A timing generation system 22 is responsive to an input clock signal 24for generating various delayed versions of the input clock signal,embodied as clock signals f_(phi0)-f_(phi6). As illustrated, these clocksignals are used to control the operation of, and more specifically theon/off timing of, the components of ADC system 10, including buffer andhold amplifiers 13-15″ and ADCs 17-17″. Bits sampled from each ADC17,17′,17″ from each stage are provided to digital error correctioncomponents such as digital error correction processor 25 for generatinga digital output signal 26 representative of input signal 11. As thebits from each ADC 17-17″ are generated at different times during theconversion process, processor 25 time-aligns all of the bits by delayingthe bits of ADC 17 and ADC 17′ such that they align with bits generatedby ADC 17″ in time. Processor 25 is configured to digitally add all ofthe bits, and output a digital representation of sampled and held inputsignal 11, which is accomplished at the output of buffer amplifier 13.

Traditional ADC clocking schemes used to generate, for example, thedelayed clock signals f_(phi0)-f_(phi6) illustrated in FIG. 1, utilizeramp-based timing in an open-loop architecture. As described above,these arrangements do not accurately track, and tend to generate highnoise. For example, FIG. 2 illustrates a timing generation system 30comprising a plurality of resistor-capacitor (RC) ramp timing generators31 ₁,31 ₂,31 _(N) and associated squaring amplifiers 32 ₁,32 ₂,32 _(N)responsive to an input clock signal 24 for generating clock signalsf_(phi0)-f_(phi6) This RC-based timing generator approach is typicallyimplemented into the above-described sub-ranging ADCs, and takesadvantage of varying RC time constants to generate the desired clockdelays. In addition to generating noise, this approach is limited tonarrow bandwidths, and does not tolerate the wide bandwidth samplingoperations required in today's more-advanced ADC applications.

ADC applications, including those used in communications, electronicwarfare, and radar systems, have diverse instantaneous bandwidth,resolution, dynamic range, and signal-to-noise requirements. FIG. 3illustrates a variation of the baseline sub-ranging ADC of FIG. 1, withlike numerals identifying like components, optimized for addressingthese multi-resolution and multi-bandwidth applications. In theillustrated approach, analog multiplexers (MUXs) 21,21′,21″ are placedin the primary analog signal path of sub-ranging ADC system 20 (i.e.between an input signal 11, through each buffer and hold amplifier13-15″, to the final ADC 17″) in order to reconfigure the system fordifferent operating resolutions. For example, if each ADC 17,17′,17″ isconfigured for five (5) bits resolution, with one (1) bit of overlap,the input signal can forego the first two of the passes and only drivethe final five (5) bit ADC 17″ directly via a single sample and holdcircuit comprising buffer amplifier 13″ and hold amplified 15″, as wellas MUXs 21,21″. Alternatively, the input signal can be multiplexed byMUX 21′ in the second pass so as to achieve nine bits of resolution,with 1 bit of overlap. Moreover, the input signal can be processed viathe primary signal path, utilizing each of the three sampling passes asdescribed above, in order to create a 13-14 bit ADC.

A drawback of this approach is that MUXs 21,21′,21″ are arrangeddirectly in the critical primary signal path of ADC system 20. Forexample, at least at the front end of ADC system 20, MUX 21 must possessa dynamic range greater than or equal to that of the converter as awhole. Further, because MUXs 21,21′,21″ are always in the primary signalpath, they will, by default, add noise, and degrade the noise, settling,and dynamic range of the converter, in comparison to the nominalconverter shown in FIG. 1. Improved converter structures must addressthese issues and accommodate multi-band and multi-mode operation, inaddition to being digitally programmable, without the above-describedlimitations.

Using the base-line architecture of sub-ranging ADC system 10 of FIG. 1,improved ADCs are realized which maintain digital programmability formulti-band, multi-mode operation, without the above-described drawbacksof sub-ranging ADC system 20 shown and described in FIG. 3. Referringgenerally to FIG. 4 a, a block diagram of a digitally reconfigurable ADCsystem 80 according to an embodiment of the present disclosure is shown.The illustrated ADC architecture places the required multiplexingfunctions in non-critical signal paths of the ADC. This results in areduction/minimization of the impact on performance for any mode ofoperation, while simultaneously providing the functionality needed toallow for a single ADC to be reused, through digital reconfiguration,for very diverse signal conditions. The ADC is digitally controlled andoffers programmable or modifiable clock rates, DC power, bandwidth,resolution, noise, linearity and distortion, as well as providing fordigital calibration and compensation of ADC gain and offset errors. Thisapproach supports moderate dynamic ranges at wide instantaneousbandwidths, as well as high dynamic range at narrow bandwidths.

Sub-ranging ADC system 80 is configured to support multi-band,multi-mode operation. For example, the ADC clock rate is variable inorder to support nearly two decades of instantaneous bandwidthrequirements (e.g. 50 MHz to 1 GHz). However, it is difficult todigitally reconfigure a single front end sample and hold circuit for theADC without compromising the performance of the sample and hold in anymode of operation. In one embodiment, the front end of ADC system 80 isconfigured for processing along two signal paths: a high dynamic range(HDR) or high resolution path 60 and high speed path 61. High resolutionsignal path 60 comprises first and second sample and hold circuits50,50′, including respective first sample and hold amplifier pair 51,53and second sample and hold amplifier pair 51′,53′. Likewise, high speedpath 61 comprises first and second sample and hold circuits 52,52′,including respective first sample and hold amplifier pair 56,57 andsecond sample and hold amplifier pair 56′,57′. Each path iscommunicatively coupled to analog input signal 11. In this way, each ofpaths 60,61 may be independently configured and optimized forperformance according to the requirements of the particular application.

Sub-ranging ADC system 80 is further configured to enable changing thedigital resolution of the converter without sacrificing performancethrough the use of MUXs arranged in the critical signal path. Due to theseparate high resolution path 60 and high speed path 61, respectivefirst sample and hold circuits 50,52, of each of these signal pathsremain separate. Depending on which configuration ADC system 80 isoperating (e.g. high speed or high resolution), the output of one ofthese first sample and hold circuits will be selectively passed to acoarse quantizer comprising ADC 87 and DAC 88 via control circuit 62 forperforming the first sampling pass. The placement of control circuit 62in this coarse path, as opposed to the main signal path of ADC system 80(e.g. between input signal 11, through each sample and hold circuit, toa last ADC 87″), relaxes the performance requirements of control circuit62, thus enabling its implementation without compromising overallperformance. In order for first sampling pass to accommodate differentpossible resolutions, the signal to ADC 87 is controlled through adigitally programmable variable attenuator 86. Additionally, the gain ofADC 87 (e.g. Q level) is digitally programmable via gain control 89.These two adjustments provide two degrees to freedom, allowing theresolution of the ADC to grow or shrink without requiring circuitmodifications.

Referring again to the main signal path of ADC system 80, in order toselect high speed path 61 or high resolution path 60, a secondmultiplexing operation is required. However, this multiplexing isaccomplished via a control circuit, including two selectively activeemitter follower paths 90,91, rather than by placing an additional MUXin the primary signal path as described above with respect to FIG. 3.This approach mitigates the noise and distortion that would result fromthe addition a MUX in the primary signal path, and enables increased ADCperformance. As illustrated in greater detail in FIG. 4 b, a controlcircuit, including parallel emitter follower transformer arrangements,is utilized to select between signals paths 60,61. In operation, thecontrol circuit acts equivalently to a MUX or a sample/hold circuit. Forselection of high speed path 61, an HS control signal is configured asbeing at a low operational level (e.g. low or “off” state) while an LScontrol signal is configured in a high operational state (high or “on”state). For the selection of the high resolution path 60 the reverse istrue. Whichever side of the illustrated differential pair is high, acurrent flows into the associated resistive loads R1,R1′, and pulls theassociated node low, thereby causing the base of the associated emitterfollower paths 90,91 to transition to a low state. Thus, the associatedtransistor Q3,Q4 will be low (off), and the other emitter followertransistor Q3,Q4 will be on (high), and the associated signal path willbe on (high). The unselected signal path is off or effectively blocked,thereby providing high isolation and low parasitic loss, so as not tocompromise the settling performance of the signal path.

Still referring to FIG. 4 b, the output of each illustrated holdamplifier 53′,57′ is connected to a base of a respective transistorQ1,Q1′ (e.g. a BJT NPN transistor). Transistors Q1,Q1′ and second diodesQ2,Q2′ (which may be embodied as a diode-connected NPN transistors)define respective emitter follower arrangements. Resistors R1,R1′provide bias current for diodes Q2,Q2′. For operation of the highresolution signal path 60, transistors Q1, Q1′, Q2, Q3 and Q5 will be on(high). While transistors Q2′, Q4 and Q6 are off (low). The controlsignal LS will be low at the base of transistor Q6, and the controlsignal HS will be high at the base of transistor Q5. Thus, for operationin the high resolution mode, transistor Q3 acts as an emitter follower,with transistors Q1,Q2 acting an up/down emitter followers,respectively. The current in the differential pair comprisingtransistors Q5,Q6 pulls the voltage at the base of transistor Q4 belowthat of transistor Q3. This causes transistor Q3 to be on, andtransistor Q4 to be off. As the current in transistor Q5 is applied toresistor R1′, the base of transistor Q4 is driven low, and the diode Q2′will also be off. Similarly, for operation of the high speed signal path61, transistors Q1, Q1′, Q2′, Q4 and Q6 are on (high), with transistorsQ2, Q3, Q5 off (low). The control signal HS will be low at the base oftransistor Q5, and the control signal LS will be high at the base oftransistor Q6. Thus, for operating in the high speed mode, transistor Q4acts as an emitter follower, with transistors Q1′,Q2′ acting an up/downemitter follower. The current in the differential pair comprisingtransistors Q5,Q6 pulls the voltage at the base of transistor Q3 belowthat of transistor Q4, causing transistor Q4 to be on and transistor Q3to be off. As the current in transistor Q6 is applied to resistor R1,the base of transistor Q3 is driven low, and the diode Q2 will also beoff.

The same topology for digitally programmable signal attenuation andquantizer gain is used in the second and third sampling stages orpasses, as illustrated. More specifically, the second pass includesamplifier 64, buffer amplifier 65, sample and hold amplifiers 66,67, asecond variable attenuator 86′ for controlling the input to signal toADC 87′ and a DAC 88′, and a gain controller 89′ for controlling gainADC 87′. The third and final sampling pass likewise includes amplifier64′, a buffer amplifier 65′, and a variable attenuator 86″ forcontrolling ADC 87″. A third digitally programmable gain controller 89″is also provided for controlling the operation of ADC 87″. Asillustrated, the resolution of each of the three sampling passes can bedigitally programmable, thereby providing flexibility to the converter'soverall resolution.

For low resolution, low bandwidth applications, the bandwidth of all thewideband circuits must be reduced. This reduction allows noise to bereduced/minimized, as well as provides for additional settling time dueto the reduced sampling rate. As discussed, it is not desirable toperform this bandwidth reduction in the critical front end highperformance sample and hold circuits of the ADC. In the illustratedembodiments of FIG. 4 a, bandwidth may be selectively reduced via theaddition of a capacitance through for example, a semiconductor basedswitch illustrated as bandwidth (BW) control circuits 70,70′,70″. Asshown in FIG. 4 a, first buffer amplifier 65 of ADC 87′ is a feedbackamplifier, and extra compensation capacitance for its control loop canbe selectively switched in to reduce the noise and bandwidth thereof.This switched in capacitor increases the settling time for the highresolution mode. After the first buffer amplifier, the performancerequirements for the second and third passes of sub-ranging ADC system80 are reduced. Thus, the capacitance of the third sample and holdcircuit can be increased in a similar fashion using additional BWcontrol circuit 70′.

In the high resolution mode of the ADC, increasing the compensationcapacitor is accomplished by turning on BW control circuit 70 (e.g. aNMOS switch) for the first summing amplifier and BW control circuits 70′for the second summing amplifier. For the high resolution mode, theinput bandwidth is reduced, as is the sample rate. Thus, for this mode,the bandwidth of the summing amplifier may be reduced, as a result ofthe increased settling time afforded to the amplifier. Additionally, alower bandwidth reduces the root mean square (RMS) noise contribution ofeach summing amplifier, thereby improving the overall signal-to-noiseratio (SNR) of the overall ADC. It should be noted that it is possibleto increase the compensation capacitor further for BW control circuit70″ compared to BW control circuit 70. This is due at least in part tothe second summing amplifier being connected further down the chain inthe sub-ranging ADC. Thus, the performance requirements for thisamplifier are reduced over the requirements for the first summingamplifier. The third sample/hold performance requirements are reducedcompared to the first two sample holds. By increasing the capacitance(C) of the hold capacitor, reductions in the sample/hold bandwidth andthe RMS noise are realized. By way of example, increasing thecapacitance by a factor of 2-4 will reduce the overall bandwidth of thesample and hold by a similar factor, as bandwidth is proportional to thevalue of

$\frac{1}{C}.$Moreover, as the root-mean-square (RMS) noise of the sample and hold isproportional to

$\frac{1}{\sqrt{C}},$the RMS noise will reduce by a factor of 1.44-2. Additionally, nonlinearpedestal and droop are reduced as the capacitance value increases. Theperformance requirements of the third pass quantizer are also reduced bythe cascaded gain of the first and second buffer amplifiers 65,65′.

While offering numerous performance advantages, sub-ranging ADC system80 of FIG. 4 a presents several challenges for implementing a suitableclock generation system. For example, the first clock signal f_(phi0)must possess very jitter for controlling the first sample and holdamplifier pair 51,53 (or 56,57). By way of non-limiting example, for arelatively high sample rate, the amount of jitter that can be toleratedin the sampling clock signal path is on the order of tens offemtoseconds. This is because the RF front end circuit of ADC system 80requires the lowest noise possible to prevent ADC performance hindrances(SNR, ENOB). For the first sample/hold in the signal chain, any jitter(noise) on the clock signal gets transferred to the held value. Afterthis event, the clock jitter requirements are relaxed, as there is nofurther sampling of the input signal. Thus, the first clock circuit mustexhibit extremely low noise. Further, timing generation requires thatedge skew outputs track over, or adjusted for, process, voltage andtemperature (PVT). More specifically, the ADC conversion process isbased on the setting of timing edges relative to each other. If theseedges drift independently of each other, then the conversion process canseverely degrade. Thus, it is important that all of the clock edgesgenerated track each other over PVT. Finally, as set forth above, thetiming generation circuit needs to cover greater than a decade offrequency coverage of the input clock frequency.

In order to support the variable frequency clock needed for the ADCapplication of FIG. 4 a, including greater than a decade of frequencycoverage, alternative clocking schemes are required. Embodiments of thepresent disclosure include clocking schemes comprising multiple delaylocked loops (DLLs), each with an octave or more of frequency coveragefor providing wide bandwidth operation (e.g. 100 MHz to 2 GHz). As thesystem is closed-loop in nature, clock outputs will by corrected forprocess and temperature.

Referring generally to FIG. 5, an exemplary single DLL 100 according toan embodiment of the present disclosure is shown. DLL 100 is responsiveto an input clock signal 24 for creating the clock edges needed for allof the functions of the ADC (e.g. ADC system 80 of FIG. 4 a). Morespecifically, an input squaring circuit 101 may be configured to performa squaring operation on input clock signal 24, and output the result(i.e. f_(phi0)) to, for example, the first sample and hold circuits ofthe ADC (e.g. sample and hold circuits 50,52 of FIG. 4 a). This approachallows for the lowest clock jitter for the first sample and hold clocksignal. In this way, aperture jitter may be reduced/minimized. This isin contrast to ADC clocking schemes using the above-described timingramps, where additional circuitry placed in-line with the input squaringcircuit may cause significant increases in clock jitter.

Still referring to FIG. 5, input clock signal 24 is provided to lownoise buffer amplifier 102, the output of which is provided to a chainof voltage controlled delay lines 103 ₁, 103 ₂, 103 ₃ . . . 103 _(N).The output of the delay line chain is provided as an input to a phasedetector 108 via a feedback delay loop 105. Phase detector 108 isconfigured to compare the phase of input clock signal 24 to the outputphase of the delay chain of DLL 100. The output of phase detector 108 isprovided to a state counter 109. As will be understood by one ofordinary skill in the art, state counter 109 is operative to determine,based on the number of delay line elements 103 ₁, 103 ₂, 103 ₃ . . . 103_(N) used in delay lock loop 100, and feedback delay loop 105, the phaseoutput of each delay element 103 ₁, 103 ₂, 103 ₃ . . . 103 _(N). In thisway, state counter 109 adjusts the phase shifts for each delay lineelement 103 ₁, 103 ₂, 103 ₃ . . . 103 _(N), such that feedback delayloop 105 will be stabilized.

The digital output of state counter 109 is provided to a phase selector111. Phase selector 111 may comprise a series of multi-bit DACs withdigital inputs provided from state counter 109. Phase selector 111 isoperative to output a plurality of control signals to respective delaylines 103 ₁, 103 ₂, 103 ₃ . . . 103 _(N) for selectively controlling thedelay imparted to input clock signal 24. The resulting delayed clocksignals output from each delay line 103 ₁, 103 ₂, 103 ₃ . . . 103 _(N)are provided to a logic circuit 110 comprising a plurality of bufferamplifiers for generating the desired remaining clock signals (e.g.f_(phi1)-f_(phi6)) for control of the subsequent sampling functions ofADC system 80 of FIG. 4 a.

Referring generally to FIG. 6, a simplified schematic diagram of a clockgeneration system according to an embodiment of the present disclosureis shown. System 92 comprises a receiver 93 responsive to input clocksignal 24. The received input clock signal 24 is provided to a pluralityof DLLs 94 ₁,94 ₂,94 _(N) (e.g. DLL 100 of FIG. 5). Each DLL 94 ₁,94₂,94 _(N) is configured to cover a portion of the exemplary 100 MHz to 2GHz operating frequency range of, for example, ADC system 80 of FIG. 4.In order to properly align the clocking edges of the output clocksignals, during test and calibration, digitally controlled DACs 96,96′can be used to adjust the placement of each clock edge 97 output from anoperating DLL via error amplifiers 95,95′.

More specifically, according to embodiments of the disclosure, clockedges 97 may be calibrated over process and temperature, and theresulting calibration data stored in look-up tables to provide forreal-time correction of timing errors in order to optimize ADCperformance. In the same fashion that DACs are used in phase selector111 to adjust phase in DLL 100 disclosed FIG. 5, DACs 96,96′ may be usedto adjust phase in the embodiment of FIG. 6. In this way, the DLL ofFIG. 5 can be considered as a coarse phase adjustment device, and DACs96,96′ of FIG. 6 can be considered as fine phase adjustment devices.

Still referring to FIG. 6, embodiments of the present disclosure includea DLL selection circuit including a multiplexer 130 and a DLL controller132 configured to provide for the selective activation and/ordeactivation of each DLL 94 ₁,94 ₂,94 _(N). Specifically, DLL controller132 may be responsive to an input control signal 134 for selectivelyactivating one (or more) of DLLs 94 ₁,94 ₂,94 _(N) via one or morecontrol signals 135 _(1-N). DLL controller 132 is further configured tooperate multiplexer 130 according to the received control signal 134 foroutputting the plurality of clock signals (e.g. f_(phi1) ⁴ _(phi6))generated by the activated or selected DLL 94 ₁,94 ₂,94 _(N). Morespecifically, multiplexer 130 may be configured to control the samplingrate to the ADC as well as the associated output clock edges for thatsample rate. DLL controller 132 provides a control signal to multiplexer130 indicative of a clock frequency range to select (e.g. one of theexemplary 3 frequency ranges). Based on this control signal multiplexer130 is configured to select the output ADC clock frequency, associatedADC clock edges, and the associated amplitude of those clock edges. Thepulse width of each ADC clock signal, for a given sample rate, isdetermined within each DLL 94 ₁,94 ₂,94 _(N). Exemplary control logic136 is shown for selectively activating a DLL according to a desiredclock frequency range.

Referring generally to FIG. 7, a clock edge control circuit or system120 for controlling DACs 96,96′ of FIG. 6 is provided. System 120 mayinclude a control processor 121 and a memory device 122 (e.g. EEROM).Memory device 122 may be used to store calibration data thereon,including pre-measured temperature calibration data associating ameasured on-chip ADC temperature with a measured clock signal edge erroror drift (i.e. a phase error). This calibration data may be stored inthe form of a look-up table (e.g. a look-up table correlating clock edgeposition or change therein to measured temperature) for each individualDAC 96,96′. In an alternate embodiment, a curve fitting function may beimplemented by control processor 121, and the appropriate edgeplacements digitally calculated for improved resolution with respect totemperature change.

One or more temperature sensors 124, such as temperature-sensing diodes,may be provided for continuously monitoring the temperature of the ADCor local junction temperature of a die in the vicinity of the DLLs.Temperature measurements may be taken periodically via processor 121,can be used to adjust timing edges in real-time based on changes intemperature during normal operation using the values in the tablelook-up. Processor 121 may be configured to receive a real-time ADCtemperature via temperature sensor 124, and associate the measuredtemperature with a known timing error stored on memory device 122.Processor 121 may be operative to provide one or more timing correctionsignals 125 based on the results of the comparison between measuredtemperature and the stored calibration data to DACs 96,96′ forcorrecting a known timing error or drift via amplifiers 95,95′. Itshould be understood that amplifiers 95,95′ provide for phaseadjustment, thereby adjusting the edges of the clock waveforms. In oneembodiment, DACs 96,96′, error amplifiers 95,95′, and control processor121 and memory device 122 may be incorporated into logic circuit 110 ofFIG. 5. In other embodiments, these components may be embodied as one ormore separate circuits.

Multiple methods may be employed to reconfigure the programmablesub-ranging ADCs described herein according to desired speed, power,resolution and noise requirements. This flexibility is generallyachieved by implementing digital programmability into select componentsof the ADC, enabling a single ADC architecture to be used in manydifferent applications. According to embodiments of the presentdisclosure, the DC current supplied to each converter element, thenumber of sub-ranging stages, the number of quantizer and DAC bits, thequantizer Q level and the DAC least significant bit (LSB) current levelall may be programmable. This is in contrast to converters in thecurrent commercial marketplace, which have no such capability to bereconfigured.

FIGS. 8 a and 8 b illustrate an exemplary system and method forreconfiguring a sub-ranging ADC having a topology similar to that shownand described with respect to FIG. 1. Sub-ranging ADC 210 comprisesbuffer and hold amplifiers (sampling gates) 213-213″, (hold amplifiers)215-215″, summing amplifiers 219,219′, ADCs 217,217′ and correspondingDACs 218,218′ making up respective first and second stages 212,214.Third stage 216 consists of a buffer amplifier 270 and ADC 217″. Adigital error correction processor 250 is provided, and operates insubstantially the same manner as described above with respect to errorcorrection processor 25 of FIG. 1. Moreover, according to an embodimentof the present disclosure, a control processor 252 is provided, and isoperative to reconfigure ADC 210 according to any desired performancecharacteristic. More specifically, ADC 210 may be configured to generatecontrol signals for selectively activating and deactivating elements offirst, second and third stages 212,214,216 in response to, for example,a user input 253.

In the exemplary illustrated configuration, a control signal 254 fromcontrol processor 252 may be used to reconfigure the digital errorcorrection logic utilized by error correction processor 250, as well asdeactivate amplifiers 213″,215″,219′, ADC 217″, DAC 218′ and all or partof ADC 217′, such that only first stage 212 and second stage 214 areoperative, while third stage 216 is bypassed. As illustrated in moredetail in FIG. 8 b, by way of example only, in response to a controlsignal generated from control processor 252, half of preamplifiers 221and comparators 222 making up ADC 217′ may be selectively deactivated,as well as a corresponding number of buffer amplifiers 223 and DACswitches 224 making up DAC 218′. While one exemplary reconfiguration isshown, it should be understood that embodiments of the presentdisclosure may operate to selectively activate and deactivate any numberand combination of components making up each and any of the ADCs andDACs of a sub-ranging ADC, and for altering the Q value and resultantresolution thereof. For example, the illustrated three stage sub-rangingADC 210 can be reduced to either one or two stages through theabove-described operations. Each of the DACs can also be reconfigured inthe same way as the ADCs. More specifically, the DAC's LSB current canbe programmed in the same way as the ADC's Q value.

In order to raise or lower the power in the preamplifiers andcomparators of the ADC and the DAC buffers, the bias current of theseindividual circuits must also be digitally programmable. However, thepreamplifiers and comparators of the ADCs (e.g. preamplifiers 221 andcomparators 222), as well as the buffer amplifiers (e.g. buffers 223) ofthe DACs, comprise resistive loads. Accordingly, as current in each ofthese circuits is reduced, for example, its output voltage, and outputvoltage swing, is also reduced. This voltage swing reduction cannegatively impact the performance of the ADC. Referring generally toFIG. 9, an exemplary circuit is shown for holding the output voltageswing of an exemplary DAC buffer 223 (e.g. a DAC buffer amplifier of DAC218′ in FIG. 8 a) constant, while the bias current in the cell isreduced, by altering the value of an internal resistive load accordingto a change in the input current. As illustrated, DAC buffer 223includes parallel nodes 270,271,272 having respective resistorsR39,R57,R24. A nominal resistance provided by resistor R24 can bereduced via the addition of resistors R39 and R57 by selectivelyactivating transistors TO and T37. In this way, the total resistive loadof DAC buffer 223 can be altered in order to track changes in inputcurrent, and the output voltage will remain constant. More specifically,is transistors TO and T37 are on or active, the load resistance consistsof a parallel combination of resistors R24, R39, and R57. This wouldcorrespond to the maximum bias current for DAC buffer 223. If the bufferbias current is now digitally reduced, the load resistance mustcorrespondingly increase. For example, transistor TO can be turned off,and now the load resistance becomes resistor R24 in parallel withresistor R57. If the bias current for DAC buffer 223 is digitallyreduced further, transistor T37 is turned off, and the value of the loadresistances becomes resistor R24. At each step of bias adjustment, theload resistance value is corresponding increased, such that the outputvoltage swing will remain the same. Thus, the exemplary DAC buffer 223provides three possible digitally programmable bias currents, and threecorresponding digitally programmable load resistor values. This topologymay be implemented into any of the components of the sub-ranging ADCshown that utilize resistive loads and differential pairs, including thepreamplifiers and comparators of the ADCs, as well as the bufferamplifiers of the DACs. For example, FIG. 10 illustrates exemplarysimulation results for the application of the above-described variableresistance/variable bias current topology implemented into a latch of acomparator (e.g. comparator 222 of FIG. 8 b). As an input current isreduced, and the load resistance is increased, and the rise (and falltime) of the comparator increases. Resultantly, the ADC power can bereduced digitally, while maintaining consistent signal swings in thesignal path.

FIGS. 11 a and 11 b illustrate additional circuitry of an exemplaryreconfigurable ADC or quantizer 217 according to an embodiment of thepresent disclosure. ADC 217 includes a resistive ladder 260 comprising aplurality of resistors 262. The quantizer Q level is set by the currentsupplied by a plurality of current sources 265,265′,265″,265″ into theresistor 262 (Q=I*R). Accordingly, the resolution and the Q level of ADC217 may be digitally programmable by adjusting the currents suppliedthese current sources. In order to adjust the resolution while adjustingthe Q value, a plurality of current sources 265,265′,265″,265″ andassociated switches 266 are provided for selectively supplying currentto resistive ladder 260. By way of example only, for full resolution andQ level operation of ADC 217, current sources 265 and 265″ may be turnedon and current sources 265′ and 265″ turned off via, for example, acontrol signal provided by control processor 252 of FIG. 8 a torespective switches 266. Resultantly, each of the associatedpre-amplifiers, comparators, DAC buffers and DAC switches are poweredon. In one implementation, to alter the ADC resolution and the Q level,current sources 265 and 265″ may be turned off and current sources 265′and 265″ turned on via respective switches 266. This arrangement resultsin only a portion of the pre-amplifiers, comparators, DAC buffers andDAC switches being powered on, with a remaining portion turned off.While FIG. 11 b illustrates two sets of current sources that can beused, additional sets of current sources and associated switches can beadded beyond what is shown to provide additional programmability. Forexample, a switchable current source may be provided between eachresistor 262 of resistive ladder 260, providing for individual controlof each of the pre-amplifiers, comparators, DAC buffers, and DACswitches.

FIG. 12 illustrates an exemplary reconfigurable DAC 218′ according to anembodiment of the present disclosure. The illustrated DAC structure isunary, in which each of current sources 270,270′ have the same currentvalue I. When an ADC is reconfigured as described above with respect toFIGS. 11 a and 11 b, the DAC also must be reconfigured accordingly. Aportion of the current sources 270 can be turned off (e.g. via one ormore switching elements responsive to one or more controls signalreceived from control processor 252 of FIG. 8 a), as an example, whenassociated pre-amplifiers and comparators are shut down in the ADC. Theremaining current sources 270′ are maintained in an active state. Inthis way, the gain of DAC 218′, or equivalently the value of the currentI, is also digitally programmable.

While these systems and methods for reconfiguring a sub-ranging ADC havebeen shown and described as applied to the simplified sub-ranging ADC ofFIG. 8 a and its components, it should be understood that these systemsand methods may be applied to any of the above-described embodiments ofsub-ranging ADCs (e.g. those described with respect to FIGS. 3-7), aswell as to any other type or topology or ADC without departing from thescope of the present invention.

While the foregoing invention has been described with reference to theabove-described embodiment, various modifications and changes can bemade without departing from the spirit of the invention. Accordingly,all such modifications and changes are considered to be within the scopeof the appended claims. Accordingly, the specification and the drawingsare to be regarded in an illustrative rather than a restrictive sense.The accompanying drawings that form a part hereof, show by way ofillustration, and not of limitation, specific embodiments in which thesubject matter may be practiced. The embodiments illustrated aredescribed in sufficient detail to enable those skilled in the art topractice the teachings disclosed herein. Other embodiments may beutilized and derived therefrom, such that structural and logicalsubstitutions and changes may be made without departing from the scopeof this disclosure. This Detailed Description, therefore, is not to betaken in a limiting sense, and the scope of various embodiments isdefined only by the appended claims, along with the full range ofequivalents to which such claims are entitled.

Such embodiments of the inventive subject matter may be referred toherein, individually and/or collectively, by the term “invention” merelyfor convenience and without intending to voluntarily limit the scope ofthis application to any single invention or inventive concept if morethan one is in fact disclosed. Thus, although specific embodiments havebeen illustrated and described herein, it should be appreciated that anyarrangement calculated to achieve the same purpose may be substitutedfor the specific embodiments shown. This disclosure is intended to coverany and all adaptations of variations of various embodiments.Combinations of the above embodiments, and other embodiments notspecifically described herein, will be apparent to those of skill in theart upon reviewing the above description.

What is claimed is:
 1. A reconfigurable wideband analog-to-digitalconverter (ADC) system comprising: a first converter stage including: afirst signal path including a first sample and hold circuit responsiveto an input signal for sampling the input signal at a first resolution;and a second signal path arranged parallel to the first signal path andincluding a second sample and hold circuit responsive to the inputsignal for sampling the input signal at a second resolution; and acontrol circuit arranged in series with the first and second signalpaths and configured to isolate one of the first and second signal pathsfrom an output of the control circuit in response to a control signal.2. The system of claim 1, wherein the control circuit comprises at leastone transistor responsive to the control signal for isolating one of thefirst and second signal paths from the output of the control circuit. 3.The system of claim 1, further comprising a first ADC configured togenerate a digital representation of the input signal sampled by thefirst converter stage.
 4. A reconfigurable wideband analog-to-digitalconverter (ADC) system comprising: a first converter stage including: afirst signal path including a first sample and hold circuit responsiveto an input signal for sampling the input signal at a first resolution;and a second signal path arranged parallel to the first signal path andincluding a second sample and hold circuit responsive to the inputsignal for sampling the input signal at a second resolution; and acontrol circuit arranged in series with the first and second signalpaths and configured to isolate one of the first and second signal pathsfrom an output of the control circuit in response to a control signal,wherein the control circuit comprises: a first transistor having a baseterminal connected to an output of the first signal path and an emitterterminal connected to the output of the control circuit; and a secondtransistor having a base terminal connected to an output of the secondsignal path and an emitter terminal connected to the output of thecontrol circuit.
 5. The system of claim 4, wherein the control circuitfurther comprises: a third transistor having a base responsive to afirst input control signal, and a collector terminal connected to thebase of the first transistor; and a fourth transistor having a baseresponsive to a second input control signal, and a collector terminalconnected to the base of the second transistor.
 6. The system of claim5, wherein applying the first input control signal to the base of thethird transistor is operative to isolate the first signal path from theoutput of the control circuit and pass the output of the second signalpath to the output of the control circuit.
 7. The system of claim 6,wherein applying the second input control signal to the base of thefourth transistor is operative to isolate the second signal path fromthe output of the system and pass the output of the first signal path tothe output of the control circuit.
 8. The system of claim 5, wherein thecontrol circuit further comprises: a fifth transistor having a baseconnected to the output of the first signal path and a collectorterminal connected to the collector terminal of the first transistor;and a sixth transistor having a base connected to the output of thesecond signal path and a collector terminal connected to the collectorterminal of the second transistor.
 9. The system of claim 8, wherein thecontrol circuit further comprises: a first resistor arranged between thecollector terminal of the fifth transistor and the base of the firsttransistor; and a second resistor arranged between the collectorterminal of the sixth transistor and the base of the second transistor.10. A reconfigurable wideband analog-to-digital converter (ADC) systemcomprising: a first converter stage including: a first signal pathincluding a first sample and hold circuit responsive to an input signalfor sampling the input signal at a first resolution; and a second signalpath arranged parallel to the first signal path and including a secondsample and hold circuit responsive to the input signal for sampling theinput signal at a second resolution; a second converter stage including:a third sample and hold circuit arranged in series with the first sampleand hold circuit; and a fourth sample and hold circuit arranged inseries with the second sample and hold circuit; and a control circuitarranged in series with the first and second signal paths for receivingthe outputs of the third and forth sample and hold circuits andconfigured to isolate one of the first and second signal paths from anoutput of the control circuit in response to a control signal.
 11. Thesystem of claim 10, further comprising a second ADC responsive to anoutput of the control circuit and configured to generate a digitalrepresentation of a signal sampled by the second converter stage. 12.The system of claim 11, further comprising: a third converter stageincluding a fifth sample and hold circuit in series with an output ofthe second converter stage; and a third ADC configured to generate adigital representation of a signal sampled by the fifth sample and holdcircuit.